Television signal scrambling system

ABSTRACT

A television signal transmission system comprises a headend including a pair of SAW filters having equal delays, one filter having amplitude and normalized phase versus frequency response characteristics continuously varying between about -6 db and 180 degrees at the picture carrier frequency and zero db and zero degrees at the sound carrier frequency of the television signal, the second filter having substantially flat amplitude and normalized phase responses between the picture and sound carrier frequencies. An RF video signal is applied to both filters, the output of the first filter being selected for transmission during at least a portion of the horizontal blanking intervals of the television signal with the output of the second filter otherwise being selected for transmission. The transmitted signal is received by a decoder including a bi-phase stable phase modulation detector and third and fourth SAW filters having response characteristics complementary to the headend filters. The received signal is applied to both the third and fourth SAW filters whose outputs are selectively switched to an output terminal in response to the phase modulation detector for restoring the received signal.

CROSS REFERENCE TO RELATED APPLICATION

This application is related to application Ser. No. 712,949 entitled"Television Signal Data Transmission System", in the names of Richard W.Citta, Dennis Mutzabaugh and Gary Sgrignoli, filed of even date herewithand assigned to Zenith Electronics Corporation.

BACKGROUND OF THE INVENTION

The present invention relates generally to television signaltransmission and receiving systems and, more particularly, to a systemwherein the horizontal synchronization components of a conventional NTSCtelevision signal are suppressed providing a scrambled transmissionformat particularly useful in subscription or pay-televisionapplications.

In subscription television systems programming signals are typicallytransmitted, either "over-the-air" or through a suitable cable network,in a scrambed form rendering the broadcast video information largelyunviewable when received by a conventional television receiver. In orderto unscramble the video display, each system subscriber is provided witha decoder operable for unscrambling the broadcast signals and forcoupling the unscrambled signals to a conventional television receiverfor viewing. Security is, of course, a prime consideration in the designof such systems. That is, the scrambling technique employed should be ofa nature making the unauthorized decoding or unscrambling of thebroadcasts difficult in order to deter the would-be pirate. At the sametime, the scrambling technique must accommodate reliable decoding byauthorized system subscribers.

One technique commonly used for scrambling the video display produced inresponse to a broadcast television signal is that of horizontal syncsuppression. Suppression of the horizontal synchronization components ofa broadcast television signal below most video levels will cause thedeflection circuits of a normal television receiver to behave eraticallysuch that a scrambled video image is produced on the display screen ofthe receiver. In addition, the ability of the television receiver to usethe color reference burst associated with the horizontal synchronizationsignals is degraded thereby causing inaccurate color reproduction.

Exemplary prior art sync supression systems are disclosed in U.S. Pat.Nos. 4,467,353 to Citta et al; 3,184,537 to Court et al; 3,478,166 toReiter et al; 3,530,232 to Reiter et al and 4,222,068 to Thompson. Inthese prior art systems, the horizontal synchronization components of abroadcast television signal are typically suppressed or reduced to graylevel and an additional keying or control signal is normally transmittedtogether with the television signal for controlling reconstruction orregeneration of the proper horizontal sync levels at the receiver. Forexample, in U.S. Pat. No. 3,184,537 an audio sub-carrier is amplitudemodulated with a suitable syncinsertion control signal. In other cases,horizontal sync reconstruction at the receiver is effected bytransmitting normal sync signals during the vertical interval of thetelevision signal for enabling a timing circuit to lock to thehorizontal components thereof. The timing circuit may then be used toaccurately define the horizontal blanking intervals of the upcomingfield to facilitate restoration of the horizontal synchronizationsignals in the composite baseband video signal. In U.S. Pat. No.4,467,353, sync reconstruction is facilitated by altering the phase ofall frequency components of the transmitted signal by an equal amountduring its horizontal blanking intervals. The present invention is animprovement of this technique involving increased system security.

OBJECTS OF THE INVENTION

It is a basic object of the present invention to provide improvedhorizontal sync suppression system of the type especially useful in asubscription television system.

It is a more specific object of the invention to provide a horizontalsync suppression system for scrambling a broadcast television signal ina manner so as to deter the unauthorized decoding of the scrambledbroadcast.

It is a further object of the invention to provide a horizontal syncsuppression system for scrambling a broadcast television signal whichdoes not require the transmission of additional control signals nor theuse of complex timing circuits to effect restoration of propersynchronization signals at a television receiver.

BRIEF DESCRIPTION OF THE DRAWINGS

Features of the invention which are believed to be novel are set forthwith particularity in the appended claims. The invention, together withits objects and the advantages thereof, may best be understood byreference to the following description taken in conjunction with theaccompanying drawings in which:

FIG. 1 is a block diagram of a television signal transmitter accordingto the invention;

FIGS. 2A and 2B are waveform diagrams illustrating a standard NTSC RFtelevision signal and the detected video envelope thereof;

FIGS. 3A-3B and 4A-4B are waveform diagrams illustrating the manner inwhich the NTSC signal of FIG. 2A is modified according to the inventionand the corresponding control signals produced in response thereto;

FIGS. 5A-5B and 6A-6B illustrate the amplitude and normalized phaseversus frequency response characteristics of the SAW filters utilized inthe transmitter and receiver of the invention;

FIG. 7 is a block diagram showing a decoder operable in response totransmitted data of the type represented in FIGS. 3A and 4A; and

FIGS. 8 and 9 are block diagrams of alternate embodiments of thetransmitter and decoder of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

Referring now to the drawings, FIG. 1 is a simplified block diagramillustrating a headend television signal scrambler and data encoder 10according to the present invention. It will be understood that thesignals provided by scrambler and data encoder 10 are intended to bebroadcast for transmission through a suitable medium such as a coaxialcable in a CATV system for reception by the system subscribers. Eachsystem subscriber is provided with a decoder which may be authorized forunscrambling a particular broadcast television program in response todata signals received from headend unit 10. Thus, with relation to aparticular television program, an authorized subscriber will be providedwith an unscrambled video image for viewing while all unauthorizedparties, whether system subscribers or not, will be provided with atelevision signal producing a scrambled video image which is largelyunintelligible when displayed on a conventional television receiverviewing screen.

With more particular reference to FIG. 1, a conventional NTSC compositebaseband video signal is coupled to an input terminal 12 of headend unit10 with the associated audio baseband signal being applied to an inputterminal 14. The audio baseband signal is coupled to an audio modulator16 where it is used to frequency modulate a sound intermediate frequency(IF) carrier, typically 41.25 MHz, which is then applied to one input ofa summing circuit 18. The composite baseband video signal is coupled toa video modulator 20 where it is used to amplitude modulate a video IFcarrier spaced 4.5 MHz from the sound lF carrier, i.e., 45.75 MHz. TheIF video signal developed at the output of modulator 20 is applied tothe inputs of a pair of filters which, in a preferred embodiment of theinvention, comprise surface acoustic wave (SAW) filters 22 and 24. Theoutputs of SAW filters 22 and 24 are selectively coupled by an RF switch26 to a second input of summing circuit 18. As will be explained infurther detail hereinafter, the frequency response characteristics ofSAW filters 22 and 24 together with the operation of RF switch 26provide for the implementation of the novel television signal scramblingand data encoding techniques of the invention.

More particularly, FIG. 2A illustrates a standard NTSC television signalof the type developed at the output of video modulator 20. The signalcomprises an IF carrier 30, having a substantially fixed phasecharacteristic and a zero carrier level 32, amplitude modulated by acomposite baseband video signal 34. Composite baseband video signal 34comprises a plurality of horizontal trace lines 36 defining the videoimage, the horizontal trace lines being separated by a plurality ofhorizontal blanking pulses 38. Each horizontal blanking pulse includes afront porch 40 and a back porch 42, the latter typically including a3.58 MHz reference color burst signal (not shown). Each horizontalblanking pulse, which defines a horizontal blanking interval, alsoincludes a horizontal synchronization pulse 44 between front porch 40and back porch 42. The horizontal synchronization pulses 44 are used tosynchronize the horizontal deflection circuits of a television receiverfor initiating horizontal retrace at the proper times, the viewingscreen of the receiver being blanked during such retrace intervals bythe horizontal blanking pulses 38.

According to NTSC standards, each horizontal blanking pulse 38 has aduration of about 12 microseconds with front porch 40 comprising about1.3 microseconds, horizontal sync pulse 44 about 5.0 microseconds andback porch 42 about 5.7 microseconds. In addition, due to thearrangement of the escutcheon in relation to the viewing screen of thereceiver, a non-viewable, overscan interval 46 of about 1.5 microsecondsis normally established at the beginning and end of each horizontaltrace line 36 immediately adjacent blanking pulse 38.

When received by a normal television receiver, the signal of FIG. 2A isdetected to reproduce composite baseband video signal 34 as illustratedby the waveform of FIG. 2B. This detection process is typically effectedby a diode-type envelope detector which will, as its name suggests,detect the envelope amplitude modulating the carrier signal 30regardless of its phase. The polarity of the detected signal in relationto the zero carrier level 32 is determined by the polarity of thedetecting diode in the receiver circuitry. As illustrated in FIG. 2B,detected composite baseband video signal 34 has a negative polarity withrelation to the zero carrier level 32 with black level signals beingmore negative than white level signals.

FIGS. 3A and 4A illustrate how the television signal of FIG. 2A ismodified by headend unit 10 to achieve the scrambling and data encodingeffects of the invention. In particular, it will be observed that thesignal is suppressed below most video levels and the phasecharacteristic of the carrier signal 30 is changed during an encodinginterval centered about each horizontal blanking pulse 38, the encodinginterval having a maximum duration T (see FIG. 2A) defined by the limitsof the overscan intervals 46 on either side thereof.

Thus, in FIG. 3A, the signal is suppressed in amplitude and the phase ofcarrier 30 is changed during an encoding interval T1 slightly less thanthe width of horizontal blanking pulse 38 while in FIG. 4A, amplitudesuppression and carrier phase change are effected during a widerencoding interval T2 extending about one microsecond into the respectiveoverscan intervals 46 on either side of horizontal blanking pulse 38.The amplitude suppression and carrier phase alteration imposed on thetelevision signal during the encoding intervals serves to effectivelyscramble the signal while the width modulation of the encoding intervalsprovides a facility for the in-band transmission of data.

Referring back to FIG. 1, the foregoing techniques are essentiallyimplemented by SAW filters 22 and 24 in combination with the operationof RF switch 26. SAW filter 22, which may have a delay of about 700nanoseconds, is designed to exhibit amplitude and normalized phaseversus frequency response characteristics as shown by the solid linecurves 51 and 53 in FIG. 5A. As used herein, the phrase normalized phaseversus frequency response means the phase versus frequency response of aparticular filter relative to the phase versus frequency response of atrue delay line of the same length; i.e., the difference between thephase versus frequency response characteristics of the filter and a truedelay line of the same length. It will be noted that the amplituderesponse 51 varies from about -6 db at the picture IF of 45.75 MHz andgradually approaches nearly 0 db at the sound IF frequency of 41.25 MHz,with a small negative offset at the chroma IF frequency of 42.17 MHz.Similarly, the normalized phase response 53 varies from about -180degrees at the picture IF and gradually approaches 0 degrees at thesound IF frequency with a small negative offset, preferably about 15degrees, at the chroma IF frequency. SAW filter 24, which has a delaymatched or equal to the delay of filter 22, is characterized bysubstantially flat amplitude and normalized phase versus frequencyresponses 55 and 57 at 0 db and 0 degrees respectively between thepicture and sound IF frequencies as shown in FIG. 6A. Alternatively, theflat responses 55 and 57 could be offset to selected small, non-zerovalues.

RF switch 26 is operated in response to a pulse width modulator 50 whichprovides a horizontal rate output defining the width of each encodinginterval. The system is operated with two different width encodingintervals T1 and T2 representing the complementary states of arespective data bit. Thus, the relatively wide encoding interval T2shown in FIG. 4A may represent a logic "1" data bit while the narrowerencoding interval T1 of FIG. 3A may represent a logic "0" data bit.Pulse width modulator 50, in response to horizontal and vertical ratetiming signals supplied by a timing generator 52, couples a horizontalrate pulse width modulated signal defining the wider and narrowerencoding intervals T1 and T2 to RF switch 26 in accordance with a datasignal supplied by a data processor 54. That is, if a particular bit ofthe data signal supplied by data processor 54 is at a logic "1" level, apulse defining the wider encoding interval T2 of FIG. 4A would beprovided to RF switch 26 by modulator 50 whereas a pulse defining thenarrower encoding interval T1 of FIG. 3A would be provided if the databit was at a logic "0" level.

RF switch 26 is operative for coupling only the output of SAW filter 24(FIG. 6A) to summing circuit 18 at all times except during an encodinginterval T1 or T2 as defined by the output of pulse width modulator 50.During an encoding interval T1 or T2 only the output of SAW filter 22(FIG. 5A) is coupled to the summing circuit. As a result, during eachencoding interval T1 or T2 the IF signal coupled to summing circuit 18by RF switch 26 is modified by the amplitude and normalized phase versusfrequency response characteristics of SAW filter 22 (FIG. 5A) to producethe scrambled and data encoded signals of FIGS. 3A and 4A. Inparticular, signal scrambling is effected by suppressing the amplitudeand altering the phase of the IF signal in accordance with the amplitudeand normalized phase response characteristics 51 and 53 during theencoding intervals and data transmission is effected by modulating thewidth of the encoding interval between the two values T1 and T2 inaccordance with the data signal supplied by data processor 54.

Since, as noted in FIG. 5A, both the attenuation and phase changeimposed on the IF signal by SAW filter 22 continuously vary between thepicture IF carrier (-6 db and -180 degrees respectively) and the soundIF carrier (substantially zero db and 0 degrees), different frequencycomponents of the signal will be subjected to varying degrees ofattenuation and phase shift. For example, due to the 180 degree phaseshift imposed on the signal at the picture IF carrier frequency, duringthe encoding interval the 3.58 MHz reference chroma burst will bedetected by a conventional intercarrier receiver at nearly the oppositepolarity in relation to an unencoded reference chroma burst. The use ofthis opposite polarity reference chroma burst to decode the colorinformation in the video signal will result in the production of colorswhich are the nearly complements of the transmitted chroma informationthereby further enhancing signal scrambling. Similarly, an intercarrierreceiver will couple a 4.5 MHz intercarrier sound signal having thewrong polarity to the sound circuits of the receiver during the encodingintervals resulting in the production of a distorted audio signal.

As mentioned previously, the width of the encoding intervals defined bythe output of pulse width modulator 50 are modulated between two valuesT1 and T2 in accordance with the data signal supplied to the modulatorby data processor 54. In a preferred embodiment of the invention, theencoding intervals are centered on the horizontal blanking pulses 38 andhave a first width or duration T2 extending about 0.5 microseconds intothe overscan intervals 46 on either side of the blanking pulse as shownby the phase reversal of carrier 30 at points 60 in FIG. 4A. It will beseen that this results in a pulse width of about 13 microseconds. Thephase reversal points 60 of the encoding interval T2 thereby extend intothe active video region of the television signal making them extremelydifficult to detect with conventional means. The second width orduration T1 of the encoding interval is preferably defined by a pair ofphase reversal points 62 respectively located in the front and backporches 40 and 42 of the horizontal blanking pulse as shown in FIG. 3A.In particular, the phase reversal points 62 are located about 0.5microseconds from either edge of the horizontal blanking pulse 38 suchthat the width of the encoding interval T1 is about 11 microseconds.Thus, in the preferred embodiment of the invention, the differencebetween the two widths T1 and T2 of the encoding intervals is about 2microseconds although satisfactory operation has been achieved withdifferences as small as 1 microsecond. It has been found that centeringthe encoding interval with respect to the horizontal blanking pulsehelps reduce audio buzz in the reproduced signal.

The scrambled and data encoded video IF signal (FIGS. 3A and 4A)produced at the output of RF switch 26 is combined in summing circuit 18with the IF audio signal and applied to an RF converter 64. RF converter64 converts the combined IF signal to a standard RF television frequencyfor transmission through the cable system. In this regard, it is to beunderstood that while the scrambling and data encoding techniques of theinvention have been described in relation to IF signals, the scramblingand data encoding could just as easily have been performed in connectionwith the transmitted channel frequency at the output of converter 64.Thus, as used herein, the term RF frequency is considered to includeboth the IF frequency as well as the transmitted channel frequency.

FIG. 7 illustrates a preferred embodiment of a decoder adapted tounscramble and decode the data in the transmitted signal. The decodercomprises an RF converter 66 which receives the transmitted signal andconverts it to the frequency of, for example, either channel 3 or 4.Assuming conversion to channel 3 frequency, the converted channel 3signal is coupled through an amplifier 68, whose gain is controlled by apotentiometer 70, to the inputs of a SAW filter 72, a SAW filter 74 anda bandpass filter 76. The output of bandpass filter 76 is coupledthrough a limiter 78 to the input of a phase modulation detector 80whose output controls an RF switch 82 for selectively coupling eitherthe output of SAW filter 72 or the output of SAW filter 74 to an outputline 84 for application to the subscriber's television receiver.

FIG. 5B illustrates the amplitude and normalized phase versus frequencyresponse characteristics 71 and 73 of SAW filter 72 at channel 3frequencies. It will be seen that these response characteristics are thecomplements of the corresponding characteristics 51 and 53 of SAW filter22 (FIG. 5A). Thus, the amplitude response 71 of filter 72 varies from+6 db at the channnel 3 picture carrier toward 0 db at the channel 3sound carrier in a manner complementary to the amplitude responsecharacteristic 51 of SAW filter 22. Similarly, the normalized phaseresponse 73 of filter 72 varies from a +180 degrees at the channel 3picture carrier toward zero degrees at the channel 3 sound carrier in amanner complementary to the phase response characteristic 53 of SAWfilter 22. It will be understood that SAW filter 72 may include suitablegain circuitry to achieve the depicted amplitude responsecharacteristic. The amplitude and normalized phase versus frequencyresponse characteristics 75 and 77 of SAW filter 74 are shown in FIG. 6Band will be seen to comprise flat responses at 0 db and 0 degreesrespectively between the channel 3 picture and sound carriers. Also,both SAW filters 72 and 74 have average delays matched to each other.

In view of the foregoing, it will be appreciated that the receivedsignal can be unscrambled by the decoder by operating RF switch 82 forcoupling the output of SAW filter 72 to output line 84 during eachencoding interval T1 and T2 and otherwise coupling the output of SAWfilter 74 to output line 84. Due to the complementary nature of theresponse characteristics of SAW filters 22 and 72, this will result inthe restoration of the amplitude and phase of the received signal duringthe encoding intervals such that the output of RF switch 82 willcorrespond to the standard NTSC waveform illustrated in FIG. 2A.

As mentioned previously, the operation of RF switch 82 is controlled byphase modulation detector 80 which, in a preferred embodiment of theinvention comprises a bi-phase stable phase modulation detector of thetype taught in U.S. Pat. No. 4,072,909. Bandpass filter 76, which alsopreferably comprises a SAW filter, has a relatively narrow response forcoupling the channel 3 picture carrier to the input of limiter 78.Limiter 78 removes the amplitude modulation from the channel 3 picturecarrier and applies the resulting amplitude limited signal to the inputof phase modulation detector 80. Detection by phase modulation detector80 is effected by a vector multiplication process whereby the appliedcarrier signal is multiplied with a fixed phase reference signal. As aresult, the amplitude limited channel 3 picture carrier applied todetector 80 will produce a detected signal exhibiting a phase reversalduring each encoding interval T1 or T2 as represented by the"super-white" level (i.e., a level above the zero carrier axis 32)pulses 90 and 92 in FIGS. 3B and 4B. Detected pulse 92 corresponds tothe wider encoding interval T2 characterizing a received signal of thetype shown in FIG. 4A while detected pulse 90 corresponds to thenarrower encoding interval T1 characterizing a received signal of thetype shown in FIG. 3A.

The detected signals illustrated in FIGS. 3B and 4B produced at theoutput of phase modulation detector 80 are applied for controlling theoperation of RF switch 82. In particular, when the detected signal ischaracterized by a low level, the output of SAW filter 74 is switched tooutput 84. However, in response to a pulse 90 or 92, the output of SAWfilter 72 is switched to output 84 for the duration of the pulse. Aspreviously explained, this operation of RF switch 82 will effectivelyunscramble the received signal.

The detected signals developed at the output of phase modulationdetector 80 are also applied to the input of a pulse width discriminator86. Pulse width discriminator 86 is responsive to the width of pulses 90and 92 for coupling complementary state logic signals to the input of adata decoder 88. For example, pulse width discriminator 86 may couple alogic "1" data bit to data decoder 88 in response to a relatiavely widepulse 92 and a logic "0" data bit in response to a narrow pulse 90. Datadecoder 88 decodes these data bits for controlling various aspects ofthe decoder. For example, the decoded data bits may represent a datamessage deauthorizing the decoder in which case an appropriate signalmay be applied over a conductor 94 for disabling RF switch 82. Ofcourse, numerous other aspects of the decoder could be controlled in asimilar manner.

FIGS. 8 and 9 illustrate alternate embodiments of the headend unit anddecoder depicted in FIGS. 1 and 7 which provide for an increased levelof signal scrambling. Referring to FIG. 8, it will be seen that theheadend unit 10 has been modified by adding a third SAW filter 96 inparallel with SAW filters 22 and 24 and by coupling an output of dataprocessor 54 to RF switch 26. SAW filter 96 has the amplitude andnormalized phase versus frequency response characteristics 91 and 93shown in dotted-line in FIG. 5A. It will be seen that the amplituderesponse 91 is similar to that of SAW filter 22 except that it has beendisplaced downwardly with the attenuation at the IF picture carrierfrequency being -10 db instead of -6 db. The normalized phase response93, on the other hand, has been folded over the zero degree axis suchthat the 180 degree phase reversal at the IF picture carrier frequencyis retained. RF switch 26 is responsive to a control signal from dataprocessor 54 for causing the switch to select either the output of SAWfilter 22 or the output of SAW filter 96 for coupling to summing circuit18 during the encoding intervals T1 and T2. The scrambling effectproduced by the selection of either filter 22 or 96 during the encodingintervals will be similar except that a greater amount of attenuationand the opposite polarity phase alteration will result when filter 96 isselected instead of filter 22.

It will be appreciated that the foregoing operation of headend unit 10results in a dynamic system having multiple scrambling modes which canbe established by selectively switching between SAW filters 22 and 96during encoding intervals T1 and T2. For example, the output of SAWfilter 22 may be coupled to summer 18 during encoding intervals T1 andT2 for a given period of time after which the output of SAW filter 96 iscoupled to summer 18 during encoding intervals T1 and T2 for anothergiven period of time, and so on. RF switch 26 couples the output ofeither SAW filter 22 or SAW filter 96 to summer 18 during encodingintervals T1 and T2 in accordance with a control signal from dataprocessor 54. In addition, data processor 54 generates and couples anappropriate mode select data message to pulse width modulator 50defining which of filters 22 or 96 has been selected. This mode selectdata message is converted by pulse width modulator 50 to a correspondingsequence of pulses 90 and 92 to effect the transmission of the datamessage to the system decoders by modulating the width of the encodingintervals as previously described. In this way, the decoders may beinstructed as to whether the output of SAW filter 22 or the output ofSAW filter 96 is being transmitted during the encoding intervals.

Referring to FIG. 9, it will be observed that the decoder has beenmodified by adding a third SAW filter 98 whose amplitude and normalizedphase versus frequency response characteristics 81 and 83 arecomplementary to those of SAW filter 96 as illustrated in dotted-line inFIG. 5B. Data decoder 88, in response to a received mode select datamessage from headend unit 10, couples a control signal to RF switch 82for selecting the output of the appropriate one of SAW filters 72 and 98during the encoding intervals for application to output 84. That is, inthe dynamic mode, data decoder 88 is responsive to received mode selectdata messages for causing the output of SAW filter 72 to be applied toconductor 84 whenever SAW filter 22 is being used at the headend and forcausing the output of SAW filter 98 to be applied to conductor 84whenever SAW filter 96 is being used at the headend.

While particular embodiments of the invention have been shown anddescribed, it will be obvious to those skilled in the art that changesand modifications may be made without departing from the invention inits broader aspects. Therefore, the aim in the appended claims is tocover all such changes and modifications as fall within the true spiritand scope of the invention.

What is claimed is:
 1. A television signal transmission system,comprising:means for developing an RF television signal including anamplitude modulated RF video component and a frequency modulated RFaudio component; means for defining a plurality of encoding intervalseach having a duration substantially time coincident with a respectivehorizontal blanking interval of said RF television signal; means forencoding said RF television signal by varying the amplitude and alteringthe phase of said video component during said encoding intervalsaccording to respective functions that vary non-linearly with frequencybetween the carrier frequencies of said video and audio components;means for transmitting said encoded RF television signal; means forreceiving said transmitted signal; phase modulation detection meansresponsive to said received signals for detecting said encodingintervals; and decoding means responsive to said detected intervals andhaving amplitude and phase response characteristics comprising thecomplements of said respect functions for restoring said televisionsignal during said encoding intervals.
 2. A television signaltransmission system according to claim 1 wherein said encoding meanscomprises means for varying the amplitude of said video componentaccording to a first non-linear function characterized by a firstpredetermined attenuation at the carrier frequency of said videocomponent and a second different attenuation at the carrier frequency ofsaid audio component and for altering the phase of said video componentaccording to a second non-linear function characterized by a firstpredetermined normalized phase shift at the carrier frequency of saidvideo component and second different predetermined normalized phaseshift at the carrier frequency of said audio component.
 3. A televisionsignal transmission system according to claim 2 wherein said means forencoding comprises a first filter having a given delay and amplitude andnormalized phase versus frequency response characteristics comprisingsaid first and second functions, a second filter having said given delayand substantially flat amplitude and normalized phase versus frequencyresponse characteristics, means for coupling said RF video component tothe inputs of said first and second filters, and means for selecting theoutput of said first filter for transmission during said encodingintervals and otherwise selecting the output of said second filter fortransmission.
 4. A television signal transmission system according toclaim 3 wherein said means for defining comprises means for definingsaid encoding intervals symmetrically about the center of saidhorizontal blanking intervals with at least some of said encodingintervals extending into the non-viewable, overscan portions of thevideo lines on either side thereof.
 5. A television signal transmissionsystem according to claim 4 wherein said decoding means comprises athird filter having a selected delay and having amplitude and normalizedphase versus frequency response characteristics comprising complementsof said first and second functions respectively, a fourth filter havingsaid selected delay and having substantially flat amplitude andnormalized phase versus frequency response characteristics, andswitching means responsive to said detected encoding intervals forcoupling the output of said third filter and otherwise coupling theoutput of said fourth filter.
 6. A television signal transmission systemaccording to claim 5 wherein said phase modulation detection meanscomprises a bi-phase stable phase modulation detector.
 7. A televisionsignal transmission system according to claim 6 wherein said first andsecond predetermined attenuations comprise a first non-zero level ofattenuation and about zero db respectively and wherein said first andsecond normalized phase shifts comprise about 180 and zero degreesrespectively.
 8. A television signal transmission system according toclaim 6 wherein said first function is characterized by a secondrelatively small non-zero level of attenuation less than said firstnon-zero level at the chroma subcarrier frequency of said RF televisionsignal and wherein said second function is characterized by a relativelysmall non-zero value of normalized phase shift less than 180 degrees atthe chroma subcarrier frequency.
 9. A television signal transmissionsystem according to claim 5 wherein said first, second, third an fourthfilters each comprise a respective SAW filter.
 10. A television signaltransmission system according to claim 8 wherein said means for encodingcomprises a fifth filter having said given delay and amplitude andnormalized phase versus frequency response characteristics that varynon-linearly with frequency between the carrier frequencies of saidvideo and audio components in a manner different from said first andsecond functions with the normalized phase shift at the video carrierfrequency being equal in magnitude and of opposite polarity to thenormalized phase shift of said first filter at the video carrierfrequency, means for coupling said RF video signal to the input of saidfifth filter, and means for selectively selecting the output of saidfifth filter or the output of said first filter for transmission duringsaid encoding intervals.
 11. A television signal transmission systemaccording to claim 10 wherein said decoding means comprises a sixthfilter having said selected delay and having amplitude and normalizedphase versus frequency response characteristics comprising thecomplements of the response characteristics of said fifth filter, saidswitching means coupling the output of said sixth filter in response tothe transmission of the output of said fifth filter.
 12. A televisionsignal transmission system according to claim 11 wherein each of saidfilters comprises a respective SAW filter.
 13. A television signaltransmission system, comprisingmeans for developing an RF televisionsignal; a first SAW filter having a given delay and having amplitude andnormalized phase versus frequency response characteristics varying as anon-linear function of frequency between a predetermined attenuation anda first polarity 180 degree phase shift at the picture carrier frequencyand substantially zero attenuation and zero phase shift at the soundcarrier frequency of said television signal; a second SAW filter havinga delay equal to said given delay and substantially flat amplitude andnormalized phase response characteristics between said picture and soundcarrier frequencies; means for applying at least the RF video componentof said television signal to said first and second SAW filters; meansfor defining a plurality of encoding intervals each having a durationsubstantially time coincident and symmetrical with a respectivehorizontal blanking interval of said television signal; first switchingmeans for selecting the output of said first SAW filter for transmissionduring said encoding intervals and otherwise selecting the output ofsaid second SAW filter for transmission; means for transmitting theselected outputs of said first and second SAW filters; means forreceiving said transmitted signal; phase modulation detection meansresponsive to said received signal for detecting said encodingintervals; third and fourth SAW filters having equal delays and havingrespective amplitude and normalized phase versus frequency responsecharacteristics which are complements of said first and second SAWfilters; means for applying said received signal to said third andfourth SAW filters; and second switch means responsive to said detectedencoding intervals for coupling the output to said third SAW filter andotherwise coupling the output of said fourth SAW filter.
 14. Atelevision signal transmission system according to claim 13 wherein atleast some of said encoding intervals extend into the non-viewable,overscan portions of the video lines on either side of the correspondinghorizontal blanking intervals.
 15. A television signal transmissionsystem according to claim 14 wherein said phase modulation detectionmeans comprises a bi-phase stable phase modulation detector.
 16. Atelevision signal transmission system according to claim 13 including afifth SAW filter having said given delay and having amplitude andnormalized phase versus frequency response characteristics varying as anon-linear function of frequency between a second predeterminedattenuation and a second opposite polarity 180 degree phase shift at thepicture carrier frequency and substantially zero attenuation and phaseshift at the sound carrier frequency of said television signal, meansfor coupling at least the RF video component of said television signalto the input of said fifth SAW filter, and means for selecting theoutput of said fifth SAW filter or said first SAW filter fortransmission during said encoding intervals.
 17. A television signaltransmission system according to claim 16 including a sixth SAW filterhaving amplitude and normalized phase versus frequency responsecharacteristics comprising the complements of the responsecharacteristics of said fifth SAW filter, said second switching meanscoupling the output of said sixth SAW filter in response to thetransmission of the output of said fifth SAW filter.
 18. A televisionsignal transmission system according to claim 17 wherein the amplitudeand normalized phase versus frequency response characteristic of saidfirst and fifth SAW filters at the chroma carrier frequency of saidtelevision signal are respectively offset in opposite polarities fromeach other by relatively small non-zero values.
 19. A television signaltransmission system according to claim 7 including a bandpass filter forcoupling the picture carrier to said received signal to said phasemodulation detector.
 20. A television signal transmission systemaccording to claim 19 including means coupled between said bandpassfilter and said phase modulation detector for limiting the amplitude ofsaid picture carrier.
 21. A television signal transmission systemaccording to claim 13 including a bandpass filter for coupling thepicture carrier of said received signal to said phase modulationdetection means.
 22. A television signal transmission system accordingto claim 21 including means coupled between said bandpass filter andsaid phase modulation detection means for limiting the amplitude of saidpicture carrier.
 23. A receiver for decoding an encoded RF televisionsignal having an RF video component whose amplitude and phase arenon-linearly altered during a plurality of encoding intervals accordingto respective first and second functions that vary with frequencybetween the carrier frequencies of the video and audio components ofsaid RF television signal, said plurality of encoding intervals eachhaving a duration substantially time coincident with a respectivehorizontal blanking interval of said RF television signal,comprising:means for receiving said encoded television signal; phasemodulation detection means responsive to said received signal fordetecting said encoding intervals; and decoding means responsive to saiddetected intervals and having amplitude and phase versus frequencyresponse characteristics respectively comprising the complements of saidfirst and second functions for restoring said RF video component duringsaid encoding intervals.
 24. A receiver according to claim 23 whereinsaid decoding means comprises an output terminal, a first filter havinga selected delay and having amplitude and normalized phase versusfrequency response characteristics comprising complements of said firstand second functions respectively, a second filter having said selecteddelay and having substantially flat amplitude and normalized phaseversus frequency response characteristics, means coupling said receivedsignal to the inputs of said first and second filters and switchingmeans responsive to said detected encoding intervals for coupling theoutput of said first filter to said output terminal and otherwisecoupling the output of said second filter to said output terminal.
 25. Areceiver according to claim 24 wherein said phase modulation detectionmeans comprises a bi-phase stable phase modulation detector.
 26. Areceiver according to claim 25 wherein each of said first and secondfilters comprises a respective SAW filter.
 27. A receiver for decodingan encoded RF television signal having an RF video component whoseamplitude and phase are non-linearly altered during a plurality ofencoding intervals according to respective first and second functions orrespective third and fourth functions, each of said functions having adifferent response characteristic that varies with frequency between thecarrier frequencies of the video and audio components of said RFtelevision signal, said plurality of encoding intervals each having aduration substantially time coincident with a respective horizontalblanking interval of said RF television signal, comprising:means forreceiving said encoded television signal; phase modulation detectionmeans responsive to said received signal for detecting said encodingintervals; a first filter having a selected delay and having amplitudeand phase versus frequency response characteristics comprisingcomplements of said first and second functions respectively; a secondfilter having said selected delay and having amplitude and phase versusfrequency response characteristics comprising complements of said thirdand fourth functions respectively; a third filter having said selecteddelay and having substantially flat amplitude and phase versus frequencyresponse characteristics; means for coupling said received signal to theinputs of said first, second and third filters; an output terminal; andmeans responsive to said detected encoding intervals for coupling theoutput of said first or second filter to said output terminal andotherwise coupling the output of said third filter to said outputterminal.
 28. A receiver according to claim 27 wherein said first,second and third filters each comprises a respective SAW filter.
 29. Areceiver for decoding an encoded RF television signal having an RF videocomponent whose amplitude and phase are non-linearly altered during aplurality of encoding intervals according to respective first and secondfunctions which vary between a predetermined attenuation and a 180degree phase shift at the picture carrier frequency and substantiallyzero attenuation and zero phase shift at the sound carrier frequency ofsaid RF television signal, said plurality of encoding intervals eachhaving a duration substantially time coincident with a respectivehorizontal blanking interval of said RF television signal,comprising:means for receiving said encoded television signal; phasemodulation detection means responsive to said received signal fordetecting said encoding intervals; a first SAW filter having a selecteddelay and having amplitude and phase versus frequency responsecharacteristics respectively comprising the complements of said firstand second functions; a second SAW filter having said selected delay andhaving substantially flat amplitude and phase versus frequency responsecharacteristics between said RF picture and sound carrier frequencies;means for applying said received signal to the inputs of said first andsecond SAW filters; an output terminal; and switch means responsive tosaid detected encoding intervals for coupling the output of said firstSAW filter to said output terminal and otherwise coupling the output ofsaid second SAW filter to said output terminal.
 30. A receiver accordingto claim 29 wherein said phase modulation detection means comprises abi-phase stable phase modulation detector.
 31. A receiver according toclaim 29 including a bandpass filter for coupling the picture carrier ofsaid received signal to said phase modulation detection means.
 32. Areceiver according to claim 31 including means coupled between saidbandpass filter and said phase modulation detection means for limitingthe amplitude of said picture carrier.
 33. A receiver according to claim29 wherein the amplitude and phase of said RF video component arealtered during said encoding intervals according to said first andsecond functions or according to respective third and fourth non-linearfunctions which vary in frequency between a second predeterminedattenuation and an opposite polarity 180 degree phase shift at thepicture carrier frequency and substantially zero attenuation and phaseshift at the sound carrier frequency of said RF television signal, saidreceiver comprising a third SAW filter having said selected delay andhaving amplitude and phase versus frequency response characteristicsrespectively comprising the complements of said third and fourthfunctions, said switch means coupling the output of said first SAWfilter or the output of said third SAW filter to said output terminal inresponse to said detected encoding intervals.